OPA656 datasheet(12/20 Pages) TI

12 / 20 page

OPA657

SBOS197B

www.ti.com

To choose the values for both CS and CF, two parameters and

only three equations need to be solved. The first parameter is

the target high-frequency noise gain NG2, which should be

greater than the minimum stable gain for the OPA657. Here,

a target NG2 of 10.5 will be used. The second parameter is the

desired low-frequency signal gain, which also sets the low-

frequency noise gain NG1. To simplify this discussion, we will

target a maximally flat 2nd-order low-pass Butterworth fre-

quency response (Q = 0.707). The signal gain of –2 shown in

Figure 4 will set the low frequency noise gain to

NG1 = 1 + RF/RG (= 3 in this example). Then, using only these

two gains and the GBP for the OPA657 (1600MHz), the key

frequency in the compensation can be determined as:

GBP

O =





















11 2

––

–

Physically, this Z0 (10.6MHz for the values shown above) is

set by 1/(2

π • R

F(CF + CS)) and is the frequency at which the

rising portion of the noise gain would intersect unity gain if

projected back to 0dB gain. The actual zero in the noise gain

occurs at NG1 • Z0 and the pole in the noise gain occurs at

NG2 • Z0. Since GBP is expressed in Hz, multiply Z0 by 2π

and use this to get CF by solving:

RZ NG

F O

π •

(= 2.86pF)

Finally, since CS and CF set the high-frequency noise gain,

determine CS by [Using NG2 = 10.5]:

CS = (NG2 – 1)CF

(= 27.2pF)

The resulting closed-loop bandwidth will be approximately

equal to:

GBP

–3

≅

(= 130MHz)

For the values shown in Figure 4, the f–3dB will be approximately

130MHz. This is less than that predicted by simply dividing the

GBP product by NG1. The compensation network controls the

bandwidth to a lower value while providing the full slew rate at

the output and an exceptional distortion performance due to

increased loop gain at frequencies below NG1 • Z0. The

capacitor values shown in Figure 4 are calculated for NG1 = 3

and NG2 = 10.5 with no adjustment for parasitics.

FIGURE 4. Broadband Low Gain Inverting External Com-

pensation.

500

Ω

27pF

OPA657

+5V

–5V

O = –2 • VI

2.9pF

250

Ω

Figure 5 shows the measured frequency response for the

circuit of Figure 4. This is showing the expected gain of –2

with exceptional flatness through 70MHz and a –3dB band-

width of 170MHz.

The real benefit to this compensation is to allow a high slew

rate, exceptional DC precision op amp to provide a low

overshoot, fast settling pulse response. For a 1V output step,

the 700V/

µs slew rate of the OPA657 will allow a rise time

limited edge rate (2ns for a 170Mhz bandwidth). While unity-

gain stable op amps may offer comparable bandwidths, their

lower slew rates will extend the settling time for larger steps.

For instance, the OPA656 can also provide a 150MHz gain of

–2 bandwidth implying a 2.3ns transition time. However, the

lower slew rate of this unity gain stable amplifier (290V/us) will

limit a 1V step transition to 3.5ns and delay the settling time as

the slewing transition is recovered. The combination of higher

slew rate and exceptional DC precision for the OPA657 can

yield one of the fastest, most precise, pulse amplifiers using

the circuit of Figure 4.

An added benefit to the compensation of Figure 4 is to

increase the loop gain above that achievable at comparable

gains by internally compensated amplifiers. The circuit of

Figure 4 will have lower harmonic distortion through 10Mhz

than the OPA656 operated at a gain of –2.

FIGURE 5. G = –2 Frequency Response with External

Compensation.

–3

–6

–9

–12

–15

–18

Frequency (MHz)

100

500

170MHz

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