OPA656 datasheet(11/20 Pages) TI

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OPA657

SBOS197B

www.ti.com

WIDEBAND, HIGH SENSITIVITY, TRANSIMPEDANCE

DESIGN

The high GBP and low input voltage and current noise for the

OPA657 make it an ideal wideband-transimpedance ampli-

fier for moderate to high transimpedance gains. Unity-gain

stability in the op amp is not required for application as a

transimpedance amplifier. One transimpedance design ex-

ample is shown on the front page of the data sheet. Designs

that require high bandwidth from a large area detector with

relatively high transimpedance gain will benefit from the low

input voltage noise for the OPA657. This input voltage noise

is peaked up over frequency by the diode source capaci-

tance, and can, in many cases, become the limiting factor to

input sensitivity. The key elements to the design are the

expected diode capacitance (CD) with the reverse bias volt-

age (–VB) applied, the desired transimpedance gain, RF, and

the GBP for the OPA657 (1600MHz). Figure 3 shows a

design from a 50pF source capacitance diode through a

200k

Ω transimpedance gain. With these 3 variables set (and

including the parasitic input capacitance for the OPA657

added to CD), the feedback capacitor value (CF) may be set

to control the frequency response.

This will give an approximate –3dB bandwidth set by:

GPB

R C

F D

−

The example of Figure 3 will give approximately 5MHz flat

bandwidth using the 0.2pF feedback compensation.

If the total output noise is bandlimited to a frequency less

than the feedback pole frequency, a very simple expression

for the equivalent input noise current can be derived as:

EC F









+ (

)

Where:

iEQ = Equivalent input noise current if the output noise is

bandlimited to F < 1/(2

πR

FCF).

iN = Input current noise for the op amp inverting input.

eN = Input voltage noise for the op amp.

CD = Diode capacitance.

F = Bandlimiting frequency in Hz (usually a postfilter prior

to further signal processing).

4kT = 1.6E – 21J at T = 290

°K

Evaluating this expression up to the feedback pole frequency

at 3.9MHz for the circuit of Figure 3, gives an equivalent input

noise current of 3.4pA/ Hz. This is much higher than the

1.2fA/ Hz for just the op amp itself. This result is being

dominated by the last term in the equivalent input noise

expression. It is essential in this case to use a low voltage

noise op amp like the OPA657. If lower transimpedance gain,

wider bandwidth solutions are needed, consider the bipolar

input OPA686 or OPA687. These parts offer comparable

gain bandwidth products but much lower input noise voltage

at the expense of higher input current noise.

LOW GAIN COMPENSATION

Where a low gain is desired, and inverting operation is

acceptable, a new external compensation technique may be

used to retain the full slew rate and noise benefits of the

OPA657 while maintaining the increased loop gain and the

associated improvement in distortion offered by the decom-

pensated architecture. This technique shapes the loop gain

for good stability while giving an easily controlled 2nd-order

low-pass frequency response. Considering only the noise

gain for the circuit of Figure 4, the low-frequency noise gain,

(NG1) will be set by the resistor ratios while the high fre-

quency noise gain (NG2) will be set by the capacitor ratios.

The capacitor values set both the transition frequencies and

the high-frequency noise gain. If this noise gain, determined

by NG2 = 1 + CS/CF, is set to a value greater than the

recommended minimum stable gain for the op amp and the

noise gain pole, set by 1/RFCF, is placed correctly, a very well

controlled 2nd-order low-pass frequency response will result.

To achieve a maximally flat 2nd-order Butterworth frequency

response, the feedback pole should be set to:

)

(

))

ππ

GPB

Adding the common-mode and differential mode input capaci-

tance (0.7 + 4.5)pF to the 50pF diode source capacitance of

Figure 3, and targeting a 200k

Ω transimpedance gain using

the 1600MHz GBP for the OPA657 will require a feedback

pole set to 3.5MHz. This will require a total feedback capaci-

tance of 0.2pF. Typical surface-mount resistors have a para-

sitic capacitance of 0.2pF, therefore, while Figure 3 shows a

0.2pF feedback-compensation capacitor, this will actually be

the parasitic capacitance of the 200k

Ω resistor.

FIGURE 3. Wideband, Low Noise, Transimpedance Amplifier.

200k

Ω

Supply Decoupling

Not Shown

50pF

OPA657

+5V

–5V

–V

O = ID RF

0.2pF

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