LM27213

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SNVS377A – FEBRUARY 2006 – REVISED MARCH 2013

Component Selection

There are numerous tradeoffs to be made in settling on a final set of component choices and as a result the

process tends to be somewhat iterative. There’s always more than one combination of parts that will work in a

given application.

We will start with a few rule of thumb assumptions and then adjust as required to find a combination that meets

the specification requirements and is cost effective. Some of the choices can be thought of as somewhat

philosophical.

Let’s start the design by choosing an inductor and then develop the remainder of the design around that choice.

INDUCTOR SELECTION

A good place to start is by choosing an appropriate buck inductor. A decent rule of thumb is to allow the worst

case, peak to peak ripple current to be on the order of 40% to 50% of the full load output current. So, for a

design of 12A at full load, the ripple current should be in the range of 4.8A to 6A. Larger or smaller ripple

currents may well be acceptable but there are tradeoffs associated with these choices. As inductor value

increases, there is a corresponding need to increase the amount of output capacitance to handle load transients.

Conversely, as inductance is reduced, the RMS switch currents tend to rise and therefore efficiency suffers

slightly while dynamic performance is improved.

The worst case ripple current will occur at the combination of maximum input and output voltage. Let’s assume

an output voltage of 1.180V and a maximum input of 16V. This will assume operation on a wall adapter while

battery voltage may be only 12V maximum. Another assumption that must be made is the intended operating

frequency. Again there exists a tradeoff between dynamic performance and efficiency. The “sweet spot” at the

time of this writing is roughly in the range of 300kHz to 400kHz. That will in all likelihood shift positive in time as

FET technology improves. The hysteretic architecture also varies the operating frequency as a function of input

voltage with the regulator tending to run a bit slower at high input voltages. Let’s assume a 300kHz frequency at

high input line. Also, since the efficiency is of somewhat less of a concern when operating from a wall adapter

we’ll design for the high end of the ripple current range under this condition. The ripple current will be lower when

operating from a battery since the input voltage will be lower and the switching frequency will be somewhat

higher. With all that settled let’s calculate a value for L.

L = (VIN-VO)VO/(ΔI x VIN x fSW)

where

•

L is the inductor value

•

Vin is the input voltage

•

Vo is the output voltage

•

ΔI is the ripple current

•

fsw is the switching frequency

(4)

So,

L = (16V-1.18V)1.18V/(6A x 16V x 300kHz)

where

•

L = 0.60 µH

(5)

If the switching frequency is pushed up a bit the inductor value may be reduced accordingly. In general for a

12A, low voltage CPU, a value between 0.56 µH and 0.7 µH works out well. The inductor chosen should be

capable of handling the full load current continuously. It must not hard saturate under fault conditions. The

saturation specifications for most inductors indicate when the inductance has fallen off by a given percentage.

This percentage will vary by manufacturer and is not standardized. As such, it’s best to look at the published

curves of inductance vs. DC current. If the inductor maintains more than 1/3 of it’s specified no load inductance

under short circuit conditions, it will probably work just fine. There will also most likely be an RMS current rating

for the inductor as well. This relates to the heating to be expected at the rated DC current. In most processor

applications it’s safe to assume the average DC current for thermal analysis purposes will be approximately 80%

of the specified maximum load current. The inductor should be specified for at least this value of continuous

current.

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