LT8331 datasheet(13/24 Pages) AD

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LT8334

Rev. 0

For more information www.analog.com

APPLICATION CIRCUITS

The LT8334 can be configured for different topologies.

The first topology to be analyzed will be the boost con-

verter, followed by the SEPIC and inverting converters.

Boost Converter: Switch Duty Cycle

The LT8334 can be configured as a boost converter for

the applications where the converter output voltage is

higher than the input voltage. Remember that boost con-

verters are not short-circuit protected. Under a shorted

output condition, the inductor current is limited only by

the input supply capability. For applications requiring a

step-up converter that is short-circuit protected, please

refer to the Applications Information section covering

SEPIC converters.

The conversion ratio as a function of duty cycles is given

by Equation 8.

VOUT

VIN

1 − D

(8)

in continuous conduction mode (CCM).

For a boost converter operating in CCM, the duty cycle

of the main switch can be calculated based on the output

voltage (VOUT) and the input voltage (VIN). The maximum

duty cycle (DMAX) occurs when the converter has the min-

imum input voltage (Equation 9).

DMAX =

VOUT − VIN(MIN)

VOUT

(9)

Discontinuous conduction mode (DCM) provides higher

conversion ratios at a given frequency at the cost of

reduced efficiencies, higher switching currents, and lower

available output power.

Boost Converter: Maximum Output Current Capability

and Inductor Selection

For the boost topology, the maximum average inductor

current is given by Equation 10.

IL(MAX)(AVG)= IO(MAX) •

− DMAX

•

(10)

where η (< 1.0) is the converter efficiency.

Due to the current limit of its internal power switch, the

LT8334 should be used in a boost converter whose max-

imum output current (IO(MAX)) is given by Equation 11.

IO(MAX) ≤

VIN(MIN)

VOUT

• 5A

− 0.5 • ΔISW

(

) • η (11)

Minimum possible inductor value and switching fre-

quency should also be considered since they will increase

inductor ripple current ∆ISW.

The inductor ripple current ∆ISW has a direct effect on

the choice of the inductor value and the converter’s max-

imum output current capability. Choosing smaller values

of ∆ISW increases output current capability but requires

large inductances and reduces the current loop gain (the

converter will approach voltage mode). Accepting larger

values of ∆ISW provides fast transient response and

allows the use of low inductances but results in higher

input current ripple and greater core losses and reduces

output current capability. It is recommended to choose a

∆ISW of approximately 1.85A.

Given an operating input voltage range, and having cho-

sen the operating frequency and ripple current in the

inductor, the inductor value of the boost converter can

be determined with Equation 12.

L =

VIN(MIN)

ΔISW • fOSC

• DMAX

(12)

The peak inductor current is the switch current limit (max-

imum 7.8A), and the RMS inductor current is approxi-

mately equal to IL(MAX)(AVG).

Choose an inductor that can handle at least 7.8A with-

out saturating and ensure that the inductor has a low

DCR (copper wire resistance) to minimize I2R power

losses. Note that in some applications, the current han-

dling requirements of the inductor can be lower, such as

in the SEPIC topology where each inductor only carries

one-half of the total switch current. For better efficiency,

use similar valued inductors with a larger volume. Many

different sizes and shapes are available from various man-

ufacturers (see Table 2). Choose a core material that has

low losses at the programmed switching frequency, such

as a ferrite core. The final value chosen for the inductor

APPLICATIONS INFORMATION

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