AD8313

–12–

REV. B

Typically, the AD8313 will need to be matched to 50

Ω. The

input impedance of the AD8313 at 100 MHz can be read from

the Smith Chart (Figure 24) and corresponds to a resistive input

impedance of 900

Ω in parallel with a capacitance of 1.1 pF.

To make the matching process simpler, the input capacitance of

the calculation later. This allows the main calculation to be

based on a simple resistive-to-resistive match (i.e., 50

Ω to

900

Ω).

The resonant frequency is defined by the equation

ω=

1

2

1

2

ω C

IN

= 2.3

µH

(C1 • C2)

(C1 + C2)

C1

C2

AD8313

50

50

SOURCE

TEMPORARY

INDUCTANCE

Figure 34. Input Matching Example

on matching a 50

Ω source resistance to a (purely resistive) load

of 900

Ω and calculating values for C

L

C

SIN

MATCH

=

1

the input will look purely resistive at a frequency given by

f

LC

O

MATCH

=

1

2

1

π

= 100 MHz

C

RR

f

pF

MATCH

SIN

O

==

11

2

75

π

.

L

RR

f

SIN

O

1

2

=

π

= 337.6 nH

L

LL

LL

MATCH

=

+

12

12

= 294 nH

C1 and C2 can be chosen in a number of ways. First C2 can be

set to a large value such as 1000 pF, so that it appears as an RF

short. C1 would then be set equal to the calculated value of

By making C1 and C2 slightly unequal (i.e., select C2 to be

about 10% less than C1) but keeping their series value the

same, the amplitude of the signals on INHI and INLO can be

equalized so that the AD8313 is driven in a more balanced

manner. Any one of the three options detailed above can be

used as long as the combined series value of C1 and C2 (i.e.,

C1

× C2/(C1 + C2)) is equal to C

from standard values. At this point, these values need now be

installed on the board and measured for performance at 100 MHz.

Because of board and layout parasitics, the component values

from the above example had to be tuned to the final values of

Assuming a lossless matching network and noting conservation

Ω

to 900

Ω) has an associated voltage gain given by

Gain

R

R

dB

IN

S

=×

20 log

= 12.6 dB

Because the AD8313 input responds to voltage and not true

power, the voltage gain of the matching network will increase

the effective input low-end power sensitivity by this amount.

Thus, in this case, the dynamic range will be shifted down-

wards, that is, the 12.6 dB voltage gain will shift the 0 dBm to

–65 dBm input range downwards to –12.6 dBm to –77.6 dBm.

However, because of network losses this gain will not be fully

realized in practice. Reference Figures 31 and 32 for an example

of practical attainable voltage gains.

Table I shows recommended values for the inductor and capaci-

tors in Figure 32 for some selected RF frequencies along with the

associated theoretical voltage gain. These values for a reactive

match are optimal for the board layout detailed as Figure 45.

As previously discussed, a modification of the board layout will

produce networks that may not perform as specified. At 2.5 GHz, a

shunt inductor is sufficient to achieve match. Consequently, C1

and C2 are set sufficiently high that they appear as RF shorts.

Figure 33

C1

C2

Voltage

(MHz)

(pF)

(pF)

(pF)

(nH)

Gain (dB)

100

8.9

22

15

270

12.6

9

1000

270

900

1.5

3

3

8.2

9.0

1.5

1000

8.2

1900

1.5

3

3

2.2

6.2

1.5

1000

2.2

2500

Large

390

390

2.2

3.2

Figure 35 shows the voltage response of the 100 MHz matching

network; note the high attenuation at lower frequencies typical

of a high-pass network.