MAX797CSLASHD datasheet(22/32 Pages) MAXIM

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Step-Down Controllers with

Synchronous Rectifier for CPU Power

______________________________________________________________________________________

______Selecting Other Components

MOSFET Switches

The two high-current N-channel MOSFETs must be

logic-level types with guaranteed on-resistance specifi-

cations at VGS = 4.5V. Lower gate threshold specs are

better (i.e., 2V max rather than 3V max). Drain-source

breakdown voltage ratings must at least equal the max-

imum input voltage, preferably with a 20% derating fac-

tor. The best MOSFETs will have the lowest

on-resistance per nanocoulomb of gate charge.

Multiplying RDS(ON) x QG provides a meaningful figure

by which to compare various MOSFETs. Newer MOS-

FET process technologies with dense cell structures

generally give the best performance. The internal gate

drivers can tolerate >100nC total gate charge, but

70nC is a more practical upper limit to maintain best

switching times.

In high-current applications, MOSFET package power

dissipation often becomes a dominant design factor.

I2R power losses are the greatest heat contributor for

both high- and low-side MOSFETs. I2R losses are dis-

tributed between Q1 and Q2 according to duty factor

(see the equations below). Switching losses affect the

upper MOSFET only, since the Schottky rectifier clamps

the switching node before the synchronous rectifier

turns on. Gate-charge losses are dissipated by the dri-

ver- er and don’t heat the MOSFET. Ensure that both

MOSFETs are within their maximum junction tempera-

ture at high ambient temperature by calculating the

temperature rise according to package thermal-resis-

tance specifications. The worst-case dissipation for the

high-side MOSFET occurs at the minimum battery volt-

age, and the worst-case for the low-side MOSFET

occurs at the maximum battery voltage.

PD (upper FET) = ILOAD2 x RDS(ON) x DUTY

VIN x CRSS

+ VIN x ILOAD x f x

(––––––––––– +20ns)

IGATE

PD (lower FET) = ILOAD2 x RDS(ON) x (1 - DUTY)

DUTY = (VOUT + VQ2) / (VIN - VQ1)

where: On-state voltage drop VQ_ = ILOAD x RDS(ON)

CRSS = MOSFET reverse transfer capacitance

IGATE = DH driver peak output current capability

(1A typically)

20ns = DH driver inherent rise/fall time

Under output short circuit, the synchronous-rectifier

MOSFET suffers extra stress and may need to be over-

sized if a continuous DC short circuit must be tolerated.

During short circuit, Q2’s duty factor can increase to

greater than 0.9 according to:

Q2 DUTY (short circuit) = 1 - [VQ2 / (VIN(MAX) - VQ1)]

where the on-state voltage drop VQ = (120mV / RSENSE)

x RDS(ON).

Rectifier Diode D1

Rectifier D1 is a clamp that catches the negative induc-

tor swing during the 110ns dead time between turning

off the high-side MOSFET and turning on the low-side.

D1 must be a Schottky type in order to prevent the

lossy parasitic MOSFET body diode from conducting. It

is acceptable to omit D1 and let the body diode clamp

the negative inductor swing, but efficiency will drop one

or two percent as a result. Use an MBR0530 (500mA

rated) type for loads up to 1.5A, a 1N5819 type for

loads up to 3A, or a 1N5822 type for loads up to 10A.

D1’s rated reverse breakdown voltage must be at least

equal to the maximum input voltage, preferably with a

20% derating factor.

Boost-Supply Diode D2

A signal diode such as a 1N4148 works well for D2 in

most applications. If the input voltage can go below 6V,

use a small (20mA) Schottky diode for slightly improved

efficiency and dropout characteristics. Don’t use large

power diodes such as 1N5817 or 1N4001, since high

junction capacitance can cause VL to be pumped up to

excessive voltages.

Rectifier Diode D3

(Transformer Secondary Diode)

The secondary diode in coupled-inductor applications

must withstand high flyback voltages greater than 60V,

which usually rules out most Schottky rectifiers.

Common silicon rectifiers such as the 1N4001 are also

prohibited, as they are far too slow. This often makes

fast silicon rectifiers such as the MURS120 the only

choice. The flyback voltage across the rectifier is relat-

ed to the VIN-VOUT difference according to the trans-

former turns ratio:

VFLYBACK = VSEC + (VIN - VOUT) x N

where: N is the transformer turns ratio SEC/PRI

VSEC is the maximum secondary DC output voltage

VOUT is the primary (main) output voltage

Subtract the main output voltage (VOUT) from VFLYBACK

in this equation if the secondary winding is returned to

VOUT and not to ground. The diode reverse breakdown

rating must also accommodate any ringing due to leak-

age inductance. D3’s current rating should be at least

twice the DC load current on the secondary output.

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